### Categorized |Diodes, Power Devices

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Posted on 16 January 2020

# Dynamic Behavior of Freewheeling and Snubber Diodes

### Turn-On Behavior

When a diode is turned on, it has to overcome the resistance of the low-doped base. The turn-on peak voltage thus increases proportionate to the base width wB. The turn-on peak voltage becomes especially critical if a significant base width wB has to be chosen due to a high blocking voltage of more than 1200 V. For this reason, optimum turn-on behavior is achieved with punch through (PT) diodes (see "P-N Junction Diode").

Freewheeling diodes always contain recombination centers to reduce charge carrier lifetime. Recombination centers (e.g.. gold) causing an increase in base resistance are to be avoided for diodes with a high blocking voltage. Recombination centers generated by platinum diffusion, electron beam radiation, or light ions will only slightly increase the turn-on overvoltage in comparison to diodes without recombination centers.

When the diode turns into the conductive state, the voltage will initially increase to the repetitive peak forward voltage VFRM before dropping to the forward voltage level again. When the current is actively switched at a very high di/dt ratio, VFRM may reach between 200 V and 300 V for an unsuitable 1700 V diode, a level which is more than 100 times the value of VF. Turning the diode on from a blocked state will result in a far higher VFRM than turning it on from a neutral state. A low VFRM is one of the most important requirements of snubber diodes, since the snubber circuit becomes effective only after diode turn-on.

The repetitive peak forward voltage is also of importance for freewheeling diodes in IGBTs which are designed for a blocking voltage greater than 1200 V. When the IGBT is turned off, a voltage spike is generated across the parasitic inductances which still superimposes the VFRM of the freewheeling diode. The sum of both components may cause critical voltage peaks. However, this measurement is not trivial, since the inductive component and VFRM cannot be told apart in application oriented chopper circuits. Measurements can be taken on an open construction directly from the diode bonding wires. Turn-on behavior of a diode is rarely important for the total power losses, since turn-on losses only amount to a small percentage of the turn-off and forward on-state losses and are therefore negligible.

### Turn-off behavior

When turning from the conductive into the blocking state, the internal diode storage charge has to be discharged. This causes a current to flow in reverse direction in the diode. The waveform of this current characterizes the reverse recovery behavior.

Figure 1. Current and voltage characteristic of the reverse recovery process of a soft-recovery diode in a circuit

Commutation velocity di/dt (Figure 1) is determined either by the switching speed of an active switch (IGBT) or by the commutation inductance. At t0 the current reaches its zero crossing. At tw the diode starts to pick up voltage. At this instant, the pn-junction in the diode becomes free of charge carriers. This constitutes a turning point in the current flow. At tirm, the reverse current reaches its maximum. After tirm has elapsed, the current declines to the reverse current. The current characteristic depends solely on the diode. A steep decline in current is referred to as snappy recovery behavior. A slow decline in current is referred to as soft recovery behavior. The value of dir/dt determines the overvoltage present in the diode, which is why soft recovery behavior is aimed for. Reverse recovery time trr is defined as the period between t0 and the moment where the current has fallen to 20% of the maximum IRRM.

### Soft recovery behavior and switching overvoltage

Differentiating between tf and ts for trr helps to define a "soft factor" as a quantitative characteristic for recovery behavior:

Soft factor        $S= \frac{t_f}{t_s}$

The soft factor should be greater than 1 in order to minimize switching overvoltages. This definition, however, is imprecise since by this criteria the current characteristic shown in Figure 2a would be regarded as snappy, whereas the current characteristic as in Figure 2b would be considered soft. Despite s > 1, there is a steep edge in a part of the reverse current flow. A definition  that refers to the maximum dif/dt during the fall time tf would be better. For a soft recovery diode, dif/dt is within the range of di/dt for ts.

Figure 2. Current characteristic for two different possibilities of snappy reverse recovery behavior

Specifying the recovery behavior at the nominal operating point only is likewise not sufficiently meaningful. It varies as a function of different circuit parameters.

• Current: Measurements have to be taken at a current flow of less than 10% and at 200% of the specified current. This approach gives proper consideration to the fact that small currents are particularly critical for the reverse recovery behavior.
• Temperature: High temperatures are often more problematic for the recovery behavior. For certain fast diodes, however, the recovery behavior will deteriorate at room temperature or below.
• Voltage applied: Higher voltage results in poorer reverse recovery behavior.
• Rate of rise for di/dt: The dependency of di/dt varies greatly in diodes made by different manufacturers. One type of diode will become "softer" when the di/dt increases, while another will become "snappier".

The best way to characterize soft recovery behavior is to measure the turn-off overvoltage under different operating conditions (IF, Tj, VCC, di/dt). In a typical application, where the chopper is in a semiconductor module, the parasitic inductance Lsges is in the range of some 10 nH. This reduces the overvoltage generated. Due to a lack of ideal switches, the voltage applied to the IGBT will drop to a certain degree during the reverse recovery phase. The voltage measured becomes

$-V(t)= -V - L_{\sigma ges} \cdot \frac{di_R}{dt} + V_{CE} (t)$

where VCE (t) is the voltage across the IGBT at the given moment in time. In 100 A soft recovery diodes with moderate rates of rise of up to 1500 A/μs and minimum parasitic inductances, V(t) will very often be smaller than VCC at any time and no voltage spikes will occur.

Figure 3. Peak voltage during commutation in dependence of the forward current as a parameter for diode switching behavior

Figure 3 compares the overvoltage of a CAL (Controlled Axial Lifetime) diode to that of a platinum diffused diode with soft recovery behavior owing to reduced p-emitter efficiency. At rated current (75 A), the platinum diffused diode is just as soft as the CAL diode. For lower currents, however, overvoltages caused by snappy switching behavior will be present in the diode. The maximum overvoltages at 10% rated current can be over 100 V. The IGBT used will switch even lower currents more slowly, and the overvoltage will decrease. By way of contrast, CAL diodes do not display significant overvoltages under any condition. Considered from the point of view of semiconductor physics, Figure 4 shows the concentration of charge carriers in the cross section of the semiconductor material during turn-off in a snappy diode and Figure 5 depicts the same for a soft recovery diode.

Figure 4. Diffusion profile and simulation of the decline in charge carriers (hole density) in a snappy diode

Under on-state load, the n- region of the diode is flooded by > 1016cm-3 of electrons and holes; the concentration of electrons (n) and holes (p) may be assumed to be equal. During the switching operation, a charge carrier hill is formed between t2 and t4 in the n- region; at the same time n ≈ p. Charge carriers are reduced toward the cathode as a result of the electron flow and toward the anode owing to the hole flow, which appears as reverse current in the outer circuit. In the case of the snappy diode (see Figure 5), the charge carrier hill will have been consumed shortly after t4 has elapsed. Between t4 and t5 , the diode will suddenly turn from its state with charge carrier hill to a state without charge carrier hill; the reverse current will snap off.

The process in a soft recovery diode is shown in Figure 5. Throughout the entire process, a charge carrier hill which feeds the reverse current is retained. At t5, the diode will already have picked up the voltage applied. The dynamic behavior described results in a tail current.

Figure 5. Diffusion profile and simulated decline in charge carriers (hole density) in a soft recovery diode

Whether soft recovery behavior will be achieved depends on how successfully this charge carrier reduction is managed. The following measures will result in softer recovery behavior:

• The width wB in the n- region is enlarged, NPT (Non Punch Through) dimensioning is used, and a region is also integrated into the diode which cannot be reached by the field at nominal voltage. This, however, will result in a stark increase in on-state voltage or in the VF /QRR relation.
• In order to restrict the increase in wB somewhat, a two-stage n- region can be used with increased doping close to the n-n+ junction. Figure 4 and Figure 5 demonstrate how a similar effect is achieved by a flat gradient at the n-n+ junction. This measure alone, however, will not be enough to achieve soft recovery behavior. Charge carrier distribution is inverted by a low-efficiency p-emitter ( see "Emitter concept").
• An axial charge carrier lifetime profile according to the CAL concept, providing for a low charge carrier life at the pn-junction, and a longer charge carrier life at the n-n+ junction.

To ensure soft recovery behavior under any conditions, several of these measures normally have to be taken at the same time. That said, the achievements made in this respect must always be assessed with a view to the extent to which a higher on-state voltage or a higher QRR is accepted.

### Minimum turn-on time

In order to reach the "soft" switching characteristics described above, the charge carriers must be granted sufficient time to reach the state of quasi-static charge carrier distribution. This is not the case for very short conduction times.

Figure 6. a) Turn-off at high interference level (pink) at VCC =1200 V (VAK -yellow) and IF =400 A (green) at tp =0.8 μs (200 ns/Div); b) Turn-off at "normal" interference level (pink) at tp =2 μs (500 ns/Div)

Figure 6 shows switching operations during very short diode turn-on times with inductive load. The fact has been taken into account that the real turn-on time of the diode is reduced by about 1 μs since the driver short-pulse suppression and td(off) of the IGBT are subtracted from tp(off) of the IGBT. For a very short turn-on time, oscillations with a high amplitude can be detected in the the current curve (green). The interference level (pink) is only a relative measurement, taken with a conductor loop above the module. These high-frequency oscillations may influence signals and logic devices and impair safe and reliable operation. For this reason, suppressing switching signals for less than 3 μs for 1200 V IGBT and < 5 μs for 1700 V is recommended.

Figure 7. Relative interference level related to the maximum value at 200 A and 125°C for IGBT turn-off signals of different durations as a function of current and temperature (a) Tj =125°C, b) Tj =-40°C)

Owing to the lower mobility of charge carriers, this effect is particularly strong at high temperatures. At -40°C, the interference levels read were only around 50% of the values measured at 125°C. The highest interference level readings were taken at half the rated current (200 A) (Figure 7). For lower currents and turn-off signals of under 2 μs, the delay and switching times were so high that the diode was no longer capable of fully turning on.

### Switching losses

The easiest way to characterize turn-on and turn-off behavior is to use a step-down converter circuit as shown in Figure 8.

Figure 8. Reverse recovery test circuit

The IGBT T1 is turned on and off twice by means of a double pulse. The rate of rise of commutation current di/dt is set by the gate series resistor RGon. VCC is the DC link voltage. Parasitic inductances Lσ1…3 are generated in the connections between capacitors, IGBT, and diode. Figure 9 shows the IGBT control signals ("driver") and the current flow in the IGBT and diode during double-pulse operation. By turning off the IGBT, the load current in the inductance LL will be taken up by the freewheeling diode. As soon as the IGBT is turned on the next time, the diode will be commutated, and at that very moment its recovery behavior will be characterized. In addition, the IGBT takes over the reverse current of the freewheeling diode during turn-on. This process is depicted at a higher time resolution in Figure 10 for a soft recovery diode.

Figure 9. Driver control signal, IGBT, and freewheeling diode current flow in a circuit during double-pulse operation

When the IGBT conducts the peak reverse current, the IGBT voltage is still on DC-link voltage level (Figure 10a). This is the moment of maximum turn-on losses in the IGBT. The diode reverse recovery characteristic may be divided into two phases: the phase of increase up to the reverse peak current and the subsequent drop in reverse current with dir /dt.

Figure 10. Current, voltage, and power losses during turn-on of a 150 A / 1700 V IGBT (a) and diode turn-off (b) during recovery behavior measurements

The second part is the tail phase where the reverse current slowly declines to zero. A trr can no longer be reasonably defined. The tail phase causes the greatest losses in the diode, since voltage is already applied to the diode. A snappy diode without tail current generates less switching losses in the diode but also high overvoltages during turn-off. The tail phase is less harmful to the IGBT since the applied voltage has already decreased at this time.

Diode switching losses in Figure 10b are represented in the same scale as for the IGBT in Figure 10a. In application, they are low compared to the switching losses in the IGBT. For the overall power losses of both IGBT and diode, it is important to keep the peak reverse current low and to have the main part of the storage charge discharged during the tail phase. The trend towards increasingly faster switching - thus reducing the switching losses in the IGBT - results in ever increasing stresses on the diode. Depending on the type of application, it may be useful with regard to the total losses to switch more slowly than recommended in the datasheet ratings.

Switching losses largely depend on 4 parameters:

• The rate of rise of commutation current di/dt or the gate resistance of the switching IGBT (Figure 11a): switching losses tend to fall in proportion to the increase in resistance; with a very small gate resistance, the existing stray inductance will limit di/dt. The range shown corresponds to a gate series resistor of 0.5 Ω up to 8.2 Ω.
• The blocking voltage (DC link voltage VCC; Figure 11b), which builds up in the component after turn-off. The dependency can be approximately calculated using an exponent of 0.6:

• Forward current IF (Figure 12 a): The higher the current, the greater the losses. However, this dependency is not linear and can be approximated using an exponent between 0.5 and 0.6:

• Junction temperature Tj (Figure 12 a): Switching losses initially rise linear to the temperature. Only above 125°C does the increase become slightly disproportionate. Using a temperature coefficient of 0.0055…0.0065, the switching losses can be calculated as a function of the temperature.

Figure 11. Dependencies of 100 A/1200 V CAL diode switching losses; a) on di/dt (@100 A, 600 V, 150 °C); b) on the DC link voltage (@100 A, R G =1 W → 2700 A/μs, 150 °C)

Figure 12. Dependencies of switching losses of a 100 A/1200 V CAL diode; a) on IF (RG =1 Ω, 600 V, 150 °C); b) on junction temperature (100 A, 600 V, RG =1 Ω)

### Dynamic ruggedness

Apart from soft switching behavior, an equally important requirement for freewheeling diodes for a voltage of 1000 V and above is dynamic ruggedness. Figure 10b shows that almost the entire DC link voltage is taken up by the diode while it is still conducting a substantial tail current. If the IGBT is switched very steeply (low gate resistance RG), the reverse peak current and tail current will rise, at the same time causing a faster decrease in VCE at the IGBT, which is then present in the diode with a correspondingly higher dv/dt. The electric field can only spread within the depleted area. Here, extreme field intensities occur and avalanche breakdown in the semiconductor at voltages far below reverse voltage level (dynamic avalanche) are inevitable. Dynamic ruggedness is the ability of a diode to manage high rates of rise of commutation current di/dt and a high DC link voltage at the same time. An alternative to dynamic ruggedness would be to limit the di/dt of the IGBT or to limit the maximum peak reverse recovery current of the diode, which amounts to the same. This will inevitably result in higher switching losses.

While the space charge region spreads, the empty part of the n - region will have a current IR flowing through it. Electrons and holes are generated at the pn-junction by dynamic avalanche. The holes move through the highly doped p region. The electrons, however, move though the n- region, resulting in the following effective doping:

$N_{eff} = N_D + p - n_{av}$

Here, nav is the density of the electrons generated by dynamic avalanche, moving from the pn-junction through the space charge region. The electrons partly compensate the hole density, thus counteracting the avalanche effect. With small forward currents, the reverse current will also decrease and, consequently, the hole density p will decrease. However, since the switching components have a higher dv/dt at low currents, the stress caused by dynamic avalanche may be higher, especially for small currents.

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