### Categorized |Design Considerations, Drivers, Power Design, Power Devices, Selection of Components

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Posted on 18 January 2019

# Gate Current and Gate Voltage Characteristics for Drivers

The switching behavior of MOSFET and IGBT modules can be largely controlled by the gate capacitance recharge speed (i.e., in this case: gate capacitance = input capacitance CGE + CCG).

In theoretical borderline cases, the gate capacitance recharge may be controlled by resistance, voltage, or current (Figure 1).

Figure 1. Gate driving process for IGBT a) Control by resistance, b) Control by voltage, c) Control by current

The most commonly used approach is to drive the system via a gate resistor (or two separate resistors for turn-on and turn-off) according to Figure 1a, since this is the most practical way of implementing the circuit. Characteristic here is the Miller plateau in the gate-source or gate-emitter voltage (Figure 2). Both switching speed and time are adjusted by RG at a continuous supply voltage VGG; the smaller the resistance RG, the shorter the switching times. In applications using modern IGBT technologies (e.g. IGBT4), it may be observed within a defined range of the gate series resistance that, contrary to expectations, the di/dt during IGBT turn-off rises in line with the increasing RG. This aspect must be considered when dimensioning the driver circuit. The disadvantage of resistance based control is that the gate capacitance tolerances of the MOSFET or IGBT will have a direct influence on switching times and switching losses. Beyond that, the maximum values for the di/dt and du/dt during switching are not fully controllable in applications that use modern IGBT technologies.

Impressed voltage at the transistor gate driven according to Figure 1b will eliminate this influence; the switching speed of the transistor is directly determined by the defined impressed gate dv/dt. Thanks to this voltage, the gate voltage characteristic does not show a Miller plateau at all or, if so, it will be minimum only. This requires sufficient driver current and voltage capacity throughout the entire switching process. Driver topologies required for voltage controlled circuits are definitely more complex and costly in comparison to resistance based control. A possible compromise would be to combine resistance and voltage control, for example, by state dependent switching of the gate series resistors or dynamic gate control driver technologies.

Current control using a "positive" and "negative" gate current generator, as shown in Figure 1c, determines the gate charge characteristics and is comparable to resistance control with respect to gate voltage characteristics. In practice, current control is also used for controlled, one-off slow turn-off due to overcurrent or short circuit.

### Control voltage ratings

Figure 2 shows the characteristics for gate current iG and gate-emitter voltage vGE in a resistance controlled circuit.

Figure 2. Gate current and voltage characteristics during turn-on and turn-off a) Turn-on, b) Turn-off

The absolute maximum control voltage VGG for both polarities must be dimensioned according to the electrical strength of the gate isolation; in datasheets, this value is usually specified at 20V for modern power MOSFETs and IGBTs. This value may not be exceeded - not even transiently. This may mean that additional measures have to be taken during turn-off.

On the other hand, RDS(on) and VCEsat decrease as the gate voltage increases. It is therefore recommended that a positive control voltage is applied, delivering a gate voltage of

VGS = +10 V for power MOSFET modules or
VGE = +15 V for IGBT modules

during stationary on-state. Most datasheet ratings are based on these measurement parameters. For logic level MOSFETs, a positive control voltage of +5 V is sufficient.

As demonstrated in Figure 2, the gate voltage for IGBTs should be negative to the emitter potential during turn-off and OFF-state. The recommended values are -5...-8...-15 V. Throughout the entire turn-off process (even if VGE approaches VGE(th)), this will maintain a negative gate current that is high enough to result in short switching times. Another, more serious disadvantage of blocking the IGBT of a bridge circuit with VGE = 0 V will occur during the reverse recovery of the parallel inverse diode of the transistor that has been turned off because of the dvCE/dt (Figure 3).

Figure 3. Cross current in an IGBT bridge arm due to turn-on by dvCE/dt feedback of T2 a) Circuit diagram, b) Current and voltage characteristics

The high dvCE/dt of the collector-emitter voltage vCE2 during the reverse recovery di/dt of D2 will effect a displacement current iV through the gate-collector capacitance CGC2 .

$i_v = C_{GC} \cdot dv_{CE}/dt$

This displacement current will cause a voltage drop across the resistance RG (or RGE/RG). If this causes vGE to rise and exceed the threshold voltage VGE(th), T2 will be driven to its active region during the reverse recovery di/dt (cross current, additional power dissipation in T1 and T2).

Other than with IGBTs, the application of a static negative gate-source voltage during off-state is not recommended to drive power MOSFETs. Parasitic turn-on with all of the consequences described above happens within the MOSFET too; at the same time, however, it will protect the transistor/diode structure of the MOSFET, whose dv/dt resistance is limited.

The equivalent circuit of a power MOSFET demonstrates the displacement current through CDS to the base of the parasitic npn-bipolar transistor as a result of dvDS/dt. If the voltage drop at the lateral p-well resistor RW reaches threshold voltage level, the bipolar transistor will be turned on parasitically, which may lead to destruction of the MOSFET as a result of power dissipation during periodic operation.

Parasitic turn-on of the MOSFET channel at VGS = 0 V over CGD will reduce dvDS/dt during blocking state and will weaken the unwelcome effect of bipolar transistor turn-on.

Furthermore, parasitic turn-on of the MOSFET channel will reduce dv/dt at the moment of the body diode's turn-off during reverse recovery, thus avoiding failure of the diode as a consequence of its restricted dynamic ruggedness.

In field applications, MOSFET driver circuits are known which switch towards 0 V during the body diode's commutation process and apply a negative gate voltage during static off-state of the switch.

### Control current ratings, driving power

The total driving power PGavg to be delivered by the driver circuit can be determined from the gate charge QGtot

$P_{Gavg} = (V_{GG+} + \vert V_{GG-} \vert ) \cdot Q_{Gtot} \cdot f_s$ where $Q_{Gtot} = C_{ERSATZ} \cdot (V_{GG+} + \vert V_{GG-} \vert )$

Peak gate current values are calculated as follows:

$I_{GMon} = (V_{GG+} + \vert V_{GG-} \vert ) / R_{Gon}$    (Ideal)

$I_{GMoff} = (V_{GG+} + \vert V_{GG-} \vert ) / R_{Goff}$  (Ideal)

The ideal calculation method neither considers the internal resistance effective in the driver output stage, nor does it take account of the dynamically effective wave impedance resulting from the driver circuit inductance and input capacitance of the IGBT/MOSFET. The smaller the external gate series resistance, the bigger the difference between ideal and real gate peak current.

Driver power per channel is calculated as follows:

$P(V_{GG+}) = V_{GG+} \cdot Q_{Gtot} \cdot f_s$

$P(V_{GG+}) = \vert V_{GG+} \vert \cdot Q_{Gtot} \cdot f_s$

where fs = switching frequency

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