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Posted on 04 May 2019

Optimizing Control of Both the Synchronous Rectifier and Primary MOSFET in Flyback Power Supplies Improves Efficiency and Reliability

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Overcomes the limitations of Schottky diode rectifier designs without the complexity of traditional synchronous rectifier implementations

By Silvestro Fimiani, Senior Product Manager, Power Integrations

Designers of flyback power supplies have generally used Schottky diode rectification in the output stage due to its simplicity and low cost. Diode-rectifier designs have proven adequate in meeting the efficiency demands of yesteryear, particularly for low current (0.5-1 A) outputs.

New market requirement

However, as regulators take aim at the operating efficiency of small power supplies used in stand-alone charger/adapters and as bias supplies for high power applications, the impressive performance of synchronous rectification becomes very attractive. Smart phones with larger screens and much higher performance provide a great example of a device that requires an increase in power and a tightening of efficiency rules, while enjoying a phenomenal rise in popularity. Smart phone battery size has increased by more than 300 % from a typical capacity of 700-900 mAhr just a few years ago to around 3000 mAhr today. For phablets and tablets it is even higher, ranging from 6000 to 10,000 mAhr. This is driving an increase in the power supply rated current - up to 400% in some cases - from 5 watts USB (5 V, 1 A) for traditional adapters to 10-20 watts (5 V, 2-4 A) for rapid charging devices.

In addition to the higher power and current, new stringent efficiency regulations such as mandatory DOE-6 (Department of Energy - Level 6) in USA and CoC V5 Tier 2 regulation in Europe have now created a pressing need for much higher efficiency.

This combination of higher power requirement, higher performance and compact size with low external touch-temperature - while still meeting the new efficiency regulations - has challenged Schottky rectifier implementations in both performance and cost.

Schottky diode vs. SR (Synchronous Rectifier)

Schottky diodes typically have a forward voltage drop of 0.4 to 0.5 V which means that in a standard 5 V output just the Schottky diode alone can result in a power loss of up to 10 %.

Synchronous Rectification (SR) can be used to boost the efficiency and reduce the heat by eliminating the lossy Schottky diodes and replacing them with an actively controlled SR MOSFET. This is made possible by the very low resistance, RDS(on) of SR MOSFETs when conducting – down to below 10 mΩ. So the forward drop in a SR MOSFET can be just 20-40 mV for a 2-4 A output current. In high current applications, this represents a dramatic reduction in power loss from 10 % for a Schottky diode to less than 1 % for a SR FET – a 10-fold improvement. Therefore, a SR technique together with secondary side regulation control is suited to enable improvements in efficiency and thermal performance. However the complexity and cost of traditional SR has prevented its wider adoption, restricting it to complex and higher power designs.

Limitations of traditional Synchronous Rectifier (SR) alternatives

The complexity of traditional SR architecture stems from the fact that the timing control in a traditional SR FET architecture is very difficult. When comparing non-synchronous and synchronous rectifiers, it is important to understand that the synchronous rectification MOSFET doesn’t simply replace the traditional Schottky diode: complex control circuitry is also required to sense and then drive the MOSFET at the correct instant to allow current to flow only in the correct direction.

Any time that the primary side FET turns on before the secondary side FET has turned off, it will cause simultaneous conduction in both the secondary and primary circuit. This effective short-circuit across the primary transformer winding results in the dreaded “shoot-through” that will destroy the primary FET. On the other hand, once the primary FET turns off if there is a delay in turning on the secondary SR FET the result is a reduction in efficiency. So designers are faced with a difficult dilemma and a significant increase in design complexity is required to overcome these design challenges.

Traditional SR solutions deploy a separate secondary-side controller to drive the SR FET. This adds cost and complexity to the circuit, restricting its uptake due to the extra cost burden. Also, with two separate controllers these designs include a delay period, called “dead-time”, providing margin and preventing switching overlap of the primary and secondary MOSFETs (shoot-through) that can result in highly destructive cross-conduction currents. The synchronous rectification MOSFET contains an integral, parasitic body diode that operates during this dead time. Unfortunately, this body diode is also lossy and slow to turn off, so it too can contribute a 1 % to 2 % drop in efficiency. To overcome this loss in efficiency a small Schottky diode, which conducts only during the dead time, can be placed in parallel with the synchronous rectification MOSFET, ensuring that the body diode never conducts. The Schottky diode used in this way is smaller and cheaper than the part required for a diode rectification design because the average diode current is low, however an efficiency loss of >0.5 % can still be expected.

So although traditional synchronous rectification (SR) has some obvious advantages, it can be very difficult to implement because the timing of the MOSFETs turn-off signal is so critical. For optimum performance it is necessary to know exactly when the primary switch is on and off. Although the state of the MOSFET can be inferred from the secondary winding, this approach does not provide the accuracy required. If a conservative prediction is made efficiency suffers; if an overly-aggressive prediction is made, shoot-through can occur. This is challenging during normal operation but it becomes increasingly difficult to guarantee shoot-through doesn’t occur under transient conditions such as output short- circuit, start-up, AC line drop outs and load steps.

An Innovative new approach

But this is about to change with Power Integrations’ new InnoSwitch™ family of ICs. For the first time, users have a shoot-through-proof design with the simplicity of a single integrated IC (Figure 1) that completely controls both the primary and secondary FET rather than two separate primary and secondary controller ICs with opto-coupled SSR (secondary side regulation).

InnoSwitch – Single IC with integrated SR and feedback

This single IC also incorporates a very high bandwidth communication link between the primary and secondary controllers - called Fluxlink™. This high speed digital communication link is incorporated in the device package through a magnetic coupling but without any magnetic cores. The material used for the manufacture of the IC package remains the same. The secondary controller acts as the master which initiates the switching process for both the secondary and primary MOSFETs, so no prediction or inference as to the state of the two MOSFETs is required. It is shoot-through-proof because the two MOSFETs are controlled deterministically and never turned on simultaneously. Using this innovative and near instantaneous communication afforded by FluxLink technology provides the secondary controller precise control of both primary MOSFET and the secondary SR MOSFET. The system achieves almost optimum turn-on and turn-off times across the entire load range, whether the power supply is operating in discontinuous mode, continuous mode, and even under fault conditions. Therefore, the power supply is intrinsically safe and it is always working at maximum efficiency.

InnoSwitch ICs also maintain full internal galvanic isolation and are safety approved to UL1577, TÜV60950. They also meet the CQC China 5,000 meter altitude requirement for creepage (see Figure 2).

InnoSwitch ICs enable designs to meet all global safety standards

An external pin-to-pin creepage gap of over 9.65 mm is achieved using a custom surface-mount package that has been specially designed for this IC family (Figure 2).

Being a recognized safety component the InnoSwitch ICs can be placed in the primary-to-secondary isolation barrier area on the PCB, so effectively the ICs take up no useable space at all (Figure 3).

An InnoSwitch IC is placed across an isolation area

Also, the design allows for direct and simple resistor divider sensing of the power supply output voltage with excellent load transient performance and keeps the no-load power consumption below 10 mW. Direct sensing is significant as it reduces the physical volume of the output capacitors required, critical to fitting designs in ever shrinking enclosure sizes. The power supply output current measurement in an InnoSwitch IC is fully integrated inside the package, eliminating external current sense circuitry altogether. This results in higher power density, reliability and improved manufacturability.

Now synchronous rectification can be used safely and reliably in higher power chargers, even those with adaptive voltage outputs such as Qualcomm’s Quickcharge™ 2.0 and MediaTek™ PE+. The ability to deliver high currents at high efficiency also makes InnoSwitch an excellent fit for the newly announced USB-PD standard that requires support for 3 A and 5 A output load currents.

In summary, InnoSwitch ICs combine the benefit of an advanced Synchronous Rectification (SR) technique with secondary side control and communication link into a single IC to meet new market requirements for higher power, performance, density, reliability and efficiency (see Table 1).

InnoSwitch IC benefits

These benefits and, indeed, the use of InnoSwitch ICs are not limited to cell phone adapters. This new powerful architecture can also be used in any application that demands greater efficiency with higher secondary currents (>1.5 A).

Additional Information:

V, 2 A adapter – just 30 components

Smart Mobile/USB Charger Design Eliminates Opto-Couplers & Meets Latest Efficiency Standards at the End of the Cable

RDK-420, a new reference design for a 5 V, 2 A CV/CC USB charger from Power Integrations, showcases the capability of the company’s InnoSwitch™-CH family of highly integrated switcher ICs to facilitate the use of synchronous rectification techniques simply and cost-effectively, enabling safe, efficient isolated power supply design. The circuit schematic is shown in figure 4.

Primary

One side of the transformer primary is connected to the rectified DC bus, the other is connected to the integrated 650 V power MOSFET inside the InnoSwitch-CH IC (U1). A low cost RCD clamp formed by D1, R1, R14 and C1 limits the peak drain voltage due to the effects of transformer and output trace inductance. The IC is self-starting, using an internal high voltage current source to charge the BPP pin capacitor (C6) when AC is first applied.

During normal operation the primary side block is powered from an auxiliary winding on the transformer. The output of this is configured as a flyback winding, rectified and filtered (D2 and C5) and fed in the BPP pin via a current limiting resistor R4.

Output regulation is achieved using On/Off control, the number of enabled switching cycles are adjusted based on the output load. At high load most switching cycles are enabled, and at light load or no-load most cycled are disabled or skipped. Once a cycle is enabled, the power MOSFET remains on until the primary current ramps to the device current limit for the specific operating state.

Secondary

The secondary side of the InnoSwitch-CH provides output voltage sensing, output current sensing and drive to a synchronous rectifier MOSFET. The secondary of the transformer is rectified by Q1 and filtered by C10. High frequency ringing during switching transients that would otherwise create high voltage across Q1 and radiated EMI is reduced via snubber components R7 and C9.

The gate of Q1 is turned on based on the winding voltage sensed via R5 and the FWD pin of the IC. In continuous conduction mode operation the power MOSFET is turned off just prior to the secondary side commanding a new switching cycle from the primary. In discontinuous mode the MOSFET is turned off when the voltage drop across the MOSFET falls below a threshold. Secondary side control of the primary side MOSFET ensure that it is never on simultaneously with the synchronous rectification MOSFET.

The MOSFET drive signal is output on the SR/P pin

The secondary side of the IC is self-powered from either the secondary winding forward voltage or the output voltage. During CV operation the output voltage powers the device, fed into the VO pin. During CC operation, when the output voltage falls the device will power itself from the secondary winding directly. During the on-time of the primary side MOSFET the forward voltage that appears across the secondary winding is used to charge the decoupling capacitor C7 via R5 and an internal regulator. The unit enters autorestart when the sensed output voltage is lower than 3 V.

Output current is sensed internally between the IS and GND pins with a threshold of 35 mV to minimize losses. Once the internal current sense threshold is exceeded, the device adjusts the number of enabled switching cycles to maintain a fixed output current.

Below the CC threshold the device operates in constant voltage mode. The output voltage is sensed via resistor divider R8 and R9 operation with a reference voltage of 1.265 V on the FB pin when at the regulation output voltage.

Input EMI Filtering

Fuse F1 provides protection against catastrophic failure of components on the primary side. An inrush limiting thermistor (RT1) is necessary due to the low surge current rating of the rectifier diodes (D1-D4) and the relatively high value and therefore low impedance of the bulk storage capacitors C2 and C4.

Physically small diodes were selected for D1-D4 due to the limited space, specifically height from PCB to case. Capacitor C2 and C4 provide filtering of the rectified AC input and together with L1 and L2 form a p (pi) filter to attenuate differential mode EMI. A low value Y capacitor (C8) reduces common mode EMI.

 

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